Process for carrying out a non-contact remote interrogation

ABSTRACT

The invention relates to a system suitable for a remote interrogation of passive transponders using chirp signals for interrogation. The transponder preferably has an encoding unit ( 11 ), a calibrating unit ( 12 ) and a measuring unit ( 13 ) each with a plurality of parallel channels ( 11.1  to  11.5, 12.1  and  13.1  to  13.2 ). The encoding unit and the calibrating unit are preferably jointly incorporated with a common delay line ( 14 ) on the same SAW chip. The interrogation signals received in the transponder via an antenna ( 10 ) are delayed characteristically and code-specifically, in particular in the encoding and calibrating unit. Decoding in the interrogation station is preformed by discrete Fourier transformation of the response signal and subsequent evaluation of the spectrum. To correct general disturbing influences on the delay of the response signal, said signal is calibrated using a single calibrating component in the response signal. Calibration occurs by appropriate shift of the spectrum of the stored response signal. For partial correction of individual disturbing influences on the delay of the response signal components, the calibrated response signal undergoes additional correction. If further measuring response signals similar to the identifying and calibrating response signals are produced then they can, for example, be used to measure temperature by appropriate evaluation of the digitally stored response signal. The preferred combination of measuring and encoding unit enables each transponder to be calibrated individually, and consequently, for example, measurement of the absolute temperature.

RELATED APPLICATION

This application is a divisional application of our application Ser. No.09/180,409, filed Jan. 4, 1999, now allowed as U.S. Pat. No. 6,407,695,which application is an USA National Stage Application under 35 U.S.C.§371 of PCT Application No. PCT/EP97/02339, filed May 7, 1997, claimingpriority from Swiss Application No. 1157/96, filed May 7, 1996.

FIELD OF THE INVENTION

The invention relates to a process for carrying out a non-contact remoteinterrogation in a system comprising a group of mobile transponders,wherein an interrogation station emits an interrogation signal, saidinterrogation signal is converted into an information-carrying responsesignal in a transponder comprising a SAW element and then returned tothe interrogation station 1.

The invention furthermore relates to an arrangement for carrying out theprocess, a transponder for such an arrangement, a SAW element suitablefor a transponder and an algorithm for the reduction of disturbinginfluences on the propagation delay of the response signal.

BACKGROUND TO THE INVENTION

Processes of the above-mentioned kind which describe the identificationof transponders are known e.g. from U.S. Pat. Nos. 4,737,790 (Skeie etal., X-Cyte Inc.), 4,734,698 (Nysen et al., X-Cyte Inc.) and 4,096,477(Epstein et al., Northwestern University). In these processes passiveSAW transponders (so-called SAW tags) are identified by means of aninterrogation station. The transponders comprise a suitably packaged SAWelement (SAW=surface acoustic wave) consisting of a piezoelectricmaterial and suitable antennas for receiving and emittingelectromagnetic waves in the range of 905-925 MHz. The SAW elementmodifies the received interrogation signal and generates a plurality ofresponse signals each having a characteristic propagation. Two differentprocesses are used for encoding the SAW element. The U.S. Pat. No.4,096,477 uses a SAW element which does not comprise reflectors and onwhich a binary encoding is realized due to the use or omission of anoutput transducer. The response signal thus contains a different,code-specific number of signal components. This means, for example, fora code 1000 (binary) that exactly one response signal component ispresent and for the code 1111 (binary) exactly 4 response signalcomponents. A short pulse is used as the interrogation signal.

In the processes described in the X-Cyte patents, the SAW elementmodifies additionally the phases of the response signals. The number ofresponse signal components is thereby independent of the realized code;this is an advantage for decoding. The interrogation signal is aso-called chirp signal whose frequency varies in serrations in the rangeof 905-925 MHz. The SAW element comprises 16 different propagation paths(acoustic encoding channels). As a result, there are 16 differentresponse signals whose signal periods are determined for all SAWelements of the system such that they each differ by one predeterminedtime interval ΔT. The response signals expanding in the different pathsthus have a time cascading which is constant (i.e. equal for all tags).When mixing the interrogation signal with the response signals, apredetermined number of known difference frequencies is generated in theinterrogation station.

The difference frequency signals correspond to the beats between theinterrogation signals and the time-delayed response signals. They areprocessed by correspondingly tuned filters. Since in each propagationpath of the SAW element attenuation or phase shift elements areincorporated corresponding to the transponder-specific code, thetransponder-specific code information can be obtained from the phases oramplitudes of the difference frequency signals.

Temperature changes and production tolerances cause disturbingpropagation variations of the response signals. As a result, phasevariations occur which distort a decoding or make it more difficult oreven impossible. Thus, a calibration process as described in theabove-mentioned U.S. Pat. No. 4,734,698 is used in practice. In thisprocess two decoding channels in the transponder must have a uniform,transponder-unspecific code. The difference between the twocorresponding response signals is used as the reference for the moreexact phase information determination of the other response signals.

The process described above or similar processes have some of thefollowing problems. At a frequency of 2.45 GHz, the phase encoding usedis very susceptible to disturbances since already little temperaturevariations lead to great phase changes. If, for example, lithium niobate(0.7%/100° C.) is used as the substrate material for the SAW element, ata temperature change of 100° C. and a relative propagation difference of100 ns, the relative phase change of the corresponding response signalsis about 230° at 905 MHz and about 615° at 2.45 GHz. At 2.45 GHzambiguity problems arise, or the reflectors must be arranged so closetogether that resolution problems arise and/or problems concerning thepositioning of the phase elements used for encoding.

A further problem results from the described kind of calibration. Theuse of two response signals having a uniform, transponder-unspecificencoding reduces the number of the response signals usable foridentification by 2. In the described embodiment of X-Cyte, the numberof independent codes is thus reduced by the factor 4×4=16 as compared tothe same system without calibration.

A further problem results from the necessity to optimally separate theresponse signals and disturbing signals. The disturbing signals aregenerated outside the transponder (e.g. by reflections of the responsesignal at metallic objects) as well as inside the transponder (e.g.multiple reflections between transducer (converter) and reflectors inthe acoustic channels). It is known that the disturbing internalreflections can be reduced effectively if in the response signal thecomponent having the longest propagation is at least twice as long asthe shortest propagation. However, in the known realization of the SAWelement and in case there is a large number (e.g. >6) of encodingchannels, this prerequisite leads to SAW elements requiring large chipsurfaces. A further problem arises from the demand for a cost-savingproduction of the SAW element. In this connection, the size of the SAWelement is an important factor for the cost per article. The smallerthis size, the cheaper is the tag. In this connection, the known SAWtags are not satisfactory.

Further problems arise in the process for encoding the SAW elementsmentioned in U.S. Pat. No. 4,096,477 in that the number of the responsesignals to be processed is code-specific, the internal disturbingsignals (multiple reflections between the transducers) are strong andnumerous and a large number of output transducers is required (e.g. 2¹⁶codes require 16 output transducers).

A further problem is to convert the incident interrogation signals mostefficiently in response signals, i.e. with minimum losses. The betterthe interrogation signals are reflected by the reflectors, the greateris e.g. the maximum reading distance that can be achieved. Suitablereflectors for this process are e.g. described in U.S. Pat. No.4,737,790. They operate with a basic frequency of about 915 MHz. Themanufacturing of reflectors with a basic frequency of about 2.45 GHz isvery difficult with respect to production technology since the width ofthe electrode fingers is about 0.4 μm and thus leads to considerablecost disadvantages. The use of reflectors operating on the thirdharmonic and having a electrode finger width of about 0.6 μm is knownfrom the prior art e.g. K. Yamanouchi, G. Shimuzu and K. Morishita, “2.5GHz-range SAW propagation and reflection characteristics and applicationof passive electronic tag and matched filter”, Proc. IEEE UltrasonicsSymposium 1993, pp 1267-1271. The exact width of the electrode fingersparticularly depends on the substrate material used. The aboveindications relate to 128°-LiNbO₃.

A further problem results from maximizing the number of possible codes(cost reduction) and, at the same time, the strength of the responsesignals (long reading distance or safe reading-out in an environmentwith strong disturbance signals) on a given chip surface.

An effective increase in the response signal strength is achieved if asfew acoustic channels as possible are to be realized. As stated in theprior art (V. P. Plessky et al., “SAW Tags: New Ideas”, 1995 IEEEUltrasonics Symposium), an additional loss of 12 dB results if insteadof a transducer having 2 acoustic channels a solution with 4 transducersand 8 acoustic channels is realized.

In order to generate a plurality of different codes with a few or evenonly one acoustic channel, it is advantageous to put several reflectorsin the same channel, as can be taken from the prior art (e.g. L. Reindlet al., “Programmable reflectors for Saw-ID-tags”, 1993 IEEE UltrasonicSymposium). If more than one reflector are located in a channel,multiple reflections between the reflectors occur, which multiplereflections may be disturbing particularly if the described positionencoding is used. The disturbing influence can be reduced if, forexample, for a plurality of reflectors a small reflectivity is selectedor if the reflectors are placed at a great distance of each other. Bothpossibilities lead to a weakening of the response signals of thecorresponding reflectors; this is disadvantageous for the identificationsystem and reduces, for example, the maximum reading distance.

BRIEF SUMMARY OF THE INVENTION

It is an object of the invention to provide a process of theabove-mentioned kind which allows an identification which is easilyrealizable and hardly susceptible to disturbances and, additionally,allows a great encoding variety (i.e. a large encoding space) and a goodutilization of the chip surface of the SAW element.

The solution according to the invention is defined by the features ofclaim 1. By combining the position encoding with a calibration,identification can most reliably be performed in particular in case of alarge temperature range and a small chip surface. Calibration can inprinciple be replaced by any kind of measurement (in particulartemperature measurement). Even the parallel use of a calibration systemand a measuring system is possible.

The interrogation station preferably uses a chirp signal for transponderinterrogation. The response signal can be stored as a sampled, digitizedtime signal and can undergo a discrete Fourier transformation (FFT) fordecoding the identifying information. In the frequency domain thepropagation differences may be determined and processed more easily.With the realization of the SAW elements according to the invention, aneffective elimination of internal disturbing signals and a cost-savingrealization of the transponders is achieved.

Encoding leads to a characteristic time delay of at least part of theresponse signal components. It is an important feature of the processthat the number of identifying signal components is independent of theimplemented code word. It is identical for all transponders of aspecific application. The known matched filter technique is suitable forthe determination.

Exactly one response signal component (calibrating response signal) isused to measure the disturbing propagation variations of the basic delayTO and to carry out the correction of the propagations of at least theidentifying response signal components. It is an important feature ofthe new process that the temperature-dependent relative propagationvariations among the identifying response signals can be neglected.

The propagations of the identifying response signals are determined suchthat each signal lies in exactly one predetermined time window(designated A,B,C,D, . . . ). The sum of all time windows is a chain ofnon-intersecting time intervals. Each time window (enumerated 0,1,2,3, .. . ) is subdivided into predefined time slots known to the system.According to the invention the propagations of the identifying responsesignals are preset such that the signal, or the center of energy mass ofthe signal, is located in one of the predefined time slots (designatede.g. A0, i.e. in time slot 0 of time window A). The entirety of usedtime slots forms a characteristic pattern on the time axis (or in thespectrum), said pattern corresponding to the code number to be read out.

The size of the used time slots depends on the bandwidth of the system,in particular of the interrogation signal. In a preferred embodiment thebandwidth of the system is 40 MHz and the size of the time slots is atleast 25 ns. The number of time windows N_(ZF) (e.g. 8) is predeterminedby the number of encoding channels or the identifying response signals,respectively. If the number of time slots per time window is N_(ZS)(e.g. 4), the following number N of possible codes results:

N=N _(ZS) ^(N) ZF, (e.g. 4⁸=2¹⁶)  (1)

It is an important advantage of the present invention that a largenumber of codes can be generated with few transducers and reflectors. Incase two encoding channels with one reflector each are used, a total ofone transducer and two reflectors is required to realize 2¹⁶ possiblecodes, if 16 time slots are provided for any one of the two timewindows. In known processes (cf. the above-mentioned patents of thecompany X-Cyte) four transducers and eight reflectors are required forthe same number of codes.

For calibrating, a uniform encoding (family code) for all transpondersof one application is preset in a coding channel or a separate channel.Thus, all calibrating response signals of one application lie within thesame time slot of the same time window. This position is known to thesystem and can be changed well-defined for each application, e.g. byshifting the calibrating response signal into a different time slot ofthe same time window. Thus, transponder families having the same codenumber but a different family code are defined.

Determination of the actual position of the calibrating response signalcomponent in the spectrum of the response signal allows, in combinationwith the knowledge of the desired position, the approximatedetermination of the disturbing change of the basic delay TO. Thesepropagation variations are caused by temperature changes in thetransponder, by the changing air gap between the transponder and theantenna of the interrogation station, by the varying length of theantenna cable and by aging or temperature changes in the electricallines between antenna and processing unit of the interrogation station.

Contrary to the calibration known from the prior art, the processaccording to the invention uses exactly one response signal componentfor calibration. This response signal component is preferably thatcomponent having the shortest propagation. It is separated from theidentifying response signal components by an off-time slot having apredetermined duration.

For carrying out the calibration, preferably

the sampled and stored response signal (time signal) is weighted with awindow function and subsequently undergoes a discrete Fouriertransformation and

in the resulting spectrum an actual position of the calibrating responsesignal component is determined and compared with a preset desiredposition.

The deviation between actual position and desired position leads to afrequency shift Δω which can be used for calibrating the time signal.

The time signal is multiplied by a function f(t)=e^(−jΔωt), which leadsto a shift of the spectrum into a position adapted to the FFT raster,and stored as a calibrated signal.

The multiplication result is weighted with a suitable window functionand undergoes a new discrete Fourier transformation.

In the received spectrum the maxima of the samples are determined in aplurality of or in all frequency windows. Subsequently, it is testedwhether the received maximum samples have a sufficient signal-to-noiseratio. If this is the case, the corresponding code is determined. Ifthis is not the case, evaluation is stopped.

In order to enable a reliable determination of the code even in case oflong codes and strong temperature influences, the response signal may bedivided into a plurality of blocks. These blocks are processedsuccessively, wherein in each block a temperature-dependent propagationor frequency shift is compensated and the corresponding identifyingresponse signal components are evaluated.

In order to further minimize temperature influences and the like whendetermining the code contained in the identifying response signalcomponents, an additional, merely calculated correction may take place.For this, a temperature-dependent frequency shift δΔω is determined fromthe spectrum of the calibrated time signal by correlating the positionsof all samples being sufficiently close to the calibrating sample withtheir desired position. (“Sufficiently close to” means that the maximumdisturbing influences to be expected can by no means cause an essentialshift of the samples taken into consideration in this first block.) Thestored and calibrated time signal is subsequently multiplied by afunction f(t)=e^(−jδΔωt), weighted with a window function and Fouriertransformed. In this way the resulting spectrum is processedsuccessively, wherein, however, in a suitable number of not yet decodedfrequency windows the maximum samples are determined and it is testedwhether they have a sufficient signal-to-noise ratio, before thecorresponding part of the code is determined.

In a further embodiment of the invention a transponder is realized suchthat in addition to identifying and calibrating response signalcomponents also measuring response signal components are transmitted tothe interrogation station. Thus, individually addressed measuring probesand measuring probes which are read out in a non-contact manner can berealized. Particularly the effect of temperature dependency of thesignal delay, which is disturbing for the identification, canpurposefully be used for temperature measurement. The calibratingresponse signals can thereby have two functions and are used forcalibrating identifying and measuring response signals.

For performing such a measurement, the SAW element generates at leasttwo response signal components for temperature measurement at the placeof the transponder. Since the SAW element (which is in this caseadditionally used as a sensor) can be identified by its code, itsindividual characteristic values can be stored in the interrogationstation. The interrogation station can thus store characteristic valuesfor a plurality of such sensor elements so that a complete evaluation ofthe incoming response signals is always possible (e.g. determination ofthe absolute temperature instead of the temperature change only).

In a further embodiment of the invention a transponder is realized suchthat the interrogation and response signals in the transponder are leadthrough a common, propagation-increasing signal line being connectedupstream or downstream of the encoding channels. Thepropagation-increasing effect of the common delay line allows a goodseparation of the response signals from disturbing influences, inparticular from disturbing environmental reflections of theinterrogation signal. The at least partially common use of the delayline allows a decreasing space-requirement for the SAW element at aconstant number of possible codes. Thus, particularly the delay linewhich, according to the prior art, is present in each individualencoding channel of a SAW tag is replaced and a common additional delayline is purposefully introduced. Thus, the encoding channels connectedupstream or downstream can be brought to a minimum length withoutshortening the entire propagation of the interrogation or responsesignal in the transponder.

As a rule, common delay lines and encoding channels are formed on onesingle SAW chip. However, it is also possible to realize the delay lineon a separate SAW chip.

Propagation-increasing signal lines different from that for the signalcomponents suitable for calibration and/or measuring purposes can beused for the identifying signal components. It is also possible thatonly some of the encoding channels are connected to the common delayline.

In order to save chip surface, the calibration channel and one encodingchannel can partially overlap, wherein a semitransparent reflector isprovided for generating the calibrating or also the encoding responsesignal component.

In a particularly preferred embodiment, the reflectors are realized suchthat they operate on the second harmonic. The period of the reflectorstructure equals the wavelength λ of the operating frequency, e.g. 2.45GHz, and the width of the electrode fingers is about half the period,e.g. about 0.8 μm at 2.45 GHz and if 128°-LiNbO₃ is used as thesubstrate material.

The function of the reflectors is similarly good for different forms ofelectrical connections between the individual electrode fingers. Forexample, all fingers can be short-circuited (FIG. 6d) or open, orconnected as pairs of two (FIG. 6e), etc. A connection can be realizedon the ends of the fingers or between them, and also combinationsthereof are possible.

The use of the reflectors according to the invention leads toconsiderable advantages as compared to the 3rd harmonic reflectors knownfrom the prior art:

1. The width of the electrode fingers is larger, about 8/6 larger; thismakes reproducible manufacturing easier;

2. The reflection coefficient of the individual electrode fingers isessentially higher;

3. A high reflectivity can be achieved with clearly fewer electrodefingers; this is an advantage in connection with surface requirement onthe chip and manufacturing costs.

The use of the reflectors according to the invention also leads toconsiderable advantages as compared to the reflectors known from theprior art and operating on the basic frequency:

1. At 128°-LiNbO₃ and an operating frequency of 2.45 GHz the fundamentalreflectors require a period (pitch) of about 0.8 μm and an electrodefinger width of about 0.4 μm. The cost-saving manufacturing of suchstructures is difficult and, for example, outside the specification forthe i-line steppers used today. In second harmonic reflectors therequired period (pitch) doubles and the width of the electrode fingersis now about 0.8 μm. Manufacturing of second harmonic reflectors withtoday's i-line steppers does thus not cause any problems.

2. In addition, in case of a greater period (pitch) and correspondinglylarger distances, electrical breakdowns between the electrode fingers bystatic or pyroelectric charging effects are less probable.

The present invention describes a reflector which can be used in atransponder and can more easily be manufactured and/or has a higherreflectivity than is known from the prior art.

If reflectors are arranged successively in an acoustic channel, inparticular disturbing multiple reflections between the reflectors arise.A disturbing influence results in particular in combination with thedescribed kind of encoding of the transponders by means of positionencoding. This is particularly true if the reflectors are located closeto each other, e.g. closer than 1 mm, for reasons of space requirementsand in order to minimize propagation losses.

The present invention describes a new type of offset reflectors which issuitable particularly for the use in transponders. The offset reflectorsresult in particular from known reflectors by modifying them inaccordance with the invention.

In sum, the use of offset reflectors leads to the following advantages:

1. Successive offset reflectors can be located closer to each other;thus, valuable chip surface can be saved;

2. Successive offset reflectors can generate stronger response signalssince they are located closer to each other and thus fewer propagationlosses result and they can have a higher reflectivity, without thesimple round-trip reflections exerting a disturbing influence;

3. Disturbing multiple reflections which are generated by reflections atthe transducer or at the opposite reflector on the other side of thetransducer can be attenuated considerably.

In a particularly preferred embodiment second harmonic reflectors areused for the realization of offset reflectors, but all types ofreflectors can be modified in accordance with the present invention. Ifsecond harmonic offset reflectors are used, in particular for anoperating frequency of 2.45 GHz it is advantageous for the manufacturingprocess that at the end of each half an additional electric connectionof the electrode fingers is introduced, which overlap partially orcompletely in the center of the reflector.

According to the invention, a transponder layout (e.g. FIG. 6c) can berealized with offset reflectors, wherein all reflectors 50.1-7 and 51contained in the drawing are offset reflectors. As compared to a similarlayout with four transducers, eight acoustic channels and one reflectorin each channel, the present embodiment comprises only two transducers,four acoustic channels and two offset reflectors in each channel. Areflector 51 is used as a calibrating reflector, wherein it creates theshortest time delay. With the same number of, for example, five timeslots per time window (one of them being an off-time slot) and a totalof seven time windows, 4⁷=16384 possible codes result for the layoutrealized in FIG. 6c and a corresponding layout with four transducers.

If only two transducers are used, the strength of the response signalsis increased by the factor 4, since when the data carrier receives theinterrogation signal, the signal is distributed to two transducers only,and only two instead of four transducers are excited after reflection.

With the layout according to FIG. 6c the surface of the chips 54 canadditionally be almost halved with the same number of codes by changingfrom four transducers with a total of eight normal reflectors to twotransducers with eight offset reflectors.

According to the invention, there is a second possibility for realizingthe offset reflector 51, i.e. in each of all four acoustic channels52.1-4 one reflector (offset or not) is located at the same distancefrom the transducers as 51, however with clearly lower reflectivities.The advantage thereof is that a strong calibration signal is generatedby all four reflectors; however, due to the lower reflectivity of theindividual reflectors, response signal disturbance by the offsetreflectors 50.1-7 is lower than would be the case if only one offsetreflector is located in the channel 52.1. The realization of a strongcalibration response signal can also be used in cases different from thementioned one.

In practice, the arrangement according to the invention comprises aplurality of mobile transponders or SAW tags (depending on theapplication, for example several tens or several ten thousands).Moreover, interrogation stations can be provided on a plurality ofplaces (depending on the respective system demands).

An arrangement for carrying out a non-contact remote interrogationaccording to the present invention comprises an interrogation stationfor emitting an interrogation signal and for receiving and evaluating aresponse signal, at least one mobile transponder having an antenna forreceiving the interrogation signal and/or emitting the response signaland a decoding unit having a plurality of parallel encoding channels forconverting the interrogation signal into an identifying response signal.According to the invention, in front of or after the at least oneencoding unit a propagation-increasing signal line common to thecorresponding encoding channels is provided.

If delay line and encoding unit are integrated on the same SAW element,the corresponding structures can be coupled by transducers (fingerelectrode structures), 90° reflectors (e.g. coupling of two parallelchannels by two 90° reflectors aligned with respect to each other) orRMSC structures (RMSC=reversing multistrip coupler). Contrary to theembodiments comprising transducers (in which parasitic reflections ofthe signal can occur at −15 to −20 dB), the 90° reflectors and the RMSCstructures are to a large extent free of disturbing internalreflections.

It is not urgently necessary that the response signal components aregenerated by, for example, code-specifically positioned reflectors. Itcan also be the case that a plurality of encoding channels is providedwith transducers emitting on one side, wherein transducers received onthe emitting side are arranged at code-specific distances, which areconnected with the antenna (via electrical lines).

Further preferred embodiments and feature combinations can be taken fromthe following detailed description and from the complete set of claims.

BRIEF DESCRIPTION OF THE DRAWINGS

The Figures used for explaining the examples show:

FIG. 1a schematic representation of an arrangement for carrying out anon-contact remote interrogation;

FIG. 2a, b time diagrams for the explanation of the position encoding incombination with calibration and measurement;

FIG. 3 a flow diagram of the signal processing for evaluating theinformation in the response signal;

FIG. 4 a schematic representation of a transponder with a measuring,calibrating and encoding unit as well as a common delay line andtransducers emitting on one side;

FIG. 5 a schematic representation of a calibrating and encoding unitrealized with a SAW element with partially common delay line, transducercoupling and transducers emitting on one side in the encoding channels;

FIG. 6a a schematic representation of an encoding unit realized on a SAWelement with common delay line, coupling via 90° reflectors andtransducers emitting on one side in the encoding channels;

FIG. 6b a schematic representation of an encoding unit realized on a SAWelement with common delay line, coupling via RMSC structures andtransducers emitting on one side in the encoding channels;

FIG. 6c a schematic representation of a calibrating and encoding unitwith offset reflectors realized on an SAW element;

FIG. 6d a preferred embodiment of a second harmonic reflector;

FIG. 6e an embodiment of a second harmonic reflector according to theinvention;

FIG. 6f an arrangement of three offset reflectors and a transduceraccording to the invention.

Basically, in the drawings equal parts have equal reference numerals.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 shows a rough block diagram of an interrogation station 1 and atransponder 2. The interrogation station 1 constantly emitsinterrogation signals Q (e.g. chirp or pulse signals) and evaluatesreceived response signals of mobile, closely arranged transponders. Thetransponder 2 is a passive element having a (preferably integrated)antenna 10, a combination of encoding unit 11, calibrating unit 12 andmeasuring unit 13 and electrical lines. It is formed, for example, as aflat plate or card and connected to the object to be identified (or atthe place of measurement) as a mark (label).

In order to achieve a higher reading distance it is, compared to theemission of a short pulse, advantageous to use a known chirp signal(linear frequency ramp) for transponder interrogation. Such a signalcomprises a significantly higher energy, without exceeding theadmissible transmitting power. In particular, the frequency domain ismore suitable for analyzing signals corresponding to a sum of sinusoidalanalysis signals than the time domain.

The interrogation station comprises a ramp generator 3 which, in apreferred embodiment, generates the control signal for the chirp signalswith a 40 MHz bandwidth and a duration of T_(BEOB)=16 ms. The steepnessof the corresponding frequency ramp is thus 2.5 Hz per nanosecond(Hz/ns). The ramp generator 3 feeds a carrier signal generator 4 (VCO)which generates a chirp signal at a center frequency of e.g. 2.44 GHz.The resulting interrogation signal is amplified by an amplifier 5 andemitted via an antenna 6.

After an appropriately selected delay T₀ has passed, the responsesignals of the transponder 2 are received. In the demodulator 7 alow-frequency signal is generated for each component of the responsesignal by inverse mixing the response signal with the interrogationsignal, the frequency of said low-frequency signal being exactlyproportional (above-mentioned 2.5 Hz/ns) to the propagation of thecorresponding signal component. The characteristic propagations of theindividual components of the response signal are converted intoproportional frequency distances (based on a suitably selected referencefrequency) and can be evaluated in the frequency domain.

Thus, there is a clear representation between frequency domain and timedomain; this makes it possible to represent the situation in any desireddescription. It is particularly suitable for the representation of partsof the invention to select a description in the time period, wherein itis assumed that the interrogation station emits a short pulse instead ofa chirp signal and the transponder reflects a plurality ofcharacteristically delayed response signals instead of a response signalwith different frequency components.

For digital processing and decoding, the low-frequency signals aresupplied to a unit with A/D converter and DSP 8.

The code is buffered, for example, in a memory 9. In the transponder 2the interrogation signal received via the antenna 10 is supplied to acombination according to the present invention of encoding unit 11,calibrating unit 12 and measuring unit 13. Combination means in thisconnection that one unit can even be missing. In the following, thefunction of the different units is explained on the basis of examples.The complete transponder is realized such that a time interval T₀ passesbefore the first signal component R_(A) is received, said time intervalT₀ having preferably half the length of the time interval T₁ (longestpropagation of a response signal component). In this way, the usefulsignal can effectively be freed from environmental disturbances causedby reflection (external reflection).

The function of the encoding unit 11 and the signal encoding accordingto the present invention will be explained exemplarily on the basis ofFIGS. 2a, b. FIG. 2a schematically shows a possible signal constellation(the time t is indicated on the abszissa). At the time t=0, for examplean interrogation signal Q is emitted in pulses with a carrier frequencyof e.g. 2.44 GHz. The pulse duration is e.g. 25 ns. A transponder beingwithin the range receives the interrogation signal Q and converts it inits encoding unit 11 containing one or a plurality of SAW elements intoa plurality of response signals having, for example, five identifyingsignal components RA, . . . , RE.

The time interval T₂=T₁−T₀, whose length is preset for the system(which, as a rule, comprises a plurality of transponders), is dividedinto a plurality of e.g. five identifying time windows A,B, . . . . Thelength of the time windows ΔT_(ZF) is the same, for example, for allidentifying time windows, e.g. 125 ns.

All transponders of one application are realized such that in any one ofthe five identifying time windows exactly one identifying signalcomponent is located. According to the invention, the position, i.e. theexact propagation, of an identifying signal component within thecorresponding time window is determined by the encoding realized in thetransponder (position encoding) except for disturbing propagationchanges.

FIG. 2b shows the decoding of the response signal. All identifying timewindows A,B, . . . having the length ΔT_(ZF) are divided into apredetermined number of, for example, five time slots A0,A1, . . .,B0,B1, . . . each. The minimum length of the time slots ΔT_(ZS) dependson the bandwidth of the interrogation system and the resolution of twoshort signal pulses which can be derived therefrom and can still beachieved. In a preferred embodiment with a chirp signal of 40 MHz, thereis thus a minimum length of 25 ns.

According to the invention, the identifying time windows A,B, . . .contain an additional off-time slot AS,BS, . . . having the lengthΔT^(S) _(ZS). Encoding takes place such that no identifying responsesignal is located in the off-time slot. Thus, the minimum possible timedistance between two identifying signal components in neighboring timewindows can be set as requested (cf. also FIG. 2b). In the presentexample, the length of the off-time slot is the same as that of theidentifying time slots in the corresponding time windows. The achievablesignal-to-noise ratio is thus increased in the decoder 8.

In a preferred embodiment the identifying time windows A,B, . . . ,Ethus contain four identifying time slots A0, . . . ,A3 or B0, . . . ,B3,etc. and an off-time slot AS or BS, etc., and all time slots in onewindow have the same length, e.g. 25 ns. The result is a time windowlength of 5×25 ns=125 ns.

For the identification of the transponder, the interrogation stationmust determine the time positions TA, . . . ,TE of the identifyingresponse signals and must allocate them to the predetermined time slots(in this case e.g. A0, B2, . . . ) for decoding. The pattern of the thusoccupied time slots of all identifying time windows determines thetransponder-specific code.

The broken lines in FIG. 2b indicate other possible positions of signalcomponents (time slots A2, B1, . . . ) that can be present in a secondtransponder having a different code.

According to equation (1), 4⁵=1024 codes can be realized with five timewindows with four identifying time slots each. If it is necessary inpractice, the number of windows or time slots may be increased (with tenwindows with four slots each, there are already 4¹⁰=2²⁰, i.e. 20 bits).

It is not necessary that the response signal components (R_(A), . . .,R_(E)) lie completely in one time slot. It is also possible that theystand out on both sides of one time slot. At least the “center of energymass”, however, must be localized sufficiently exactly. However,overlapping reduces the signal-to-noise ratio and thus the readingdistance.

A process for decoding the information contained in the identifyingresponse signal, which process can, according to the invention, be usedfor an interrogation system with chirp signals, is described in thefollowing. The individual process steps are indicated in FIG. 3.

The transformation for processing in the frequency domain (spectrum ofthe response signal) is advantageously realized by a discrete Fouriertransformation (FFT) making use of a digital signal processor (DSP).According to the invention, the scanning frequency for thediscretization is selected such that the possible response signalsexactly meet a sample of the discrete Fourier transformation. As aresult, according to a preferred embodiment, there is a sample distanceof 62.5 Hz (=25 ns×2.5 Hz/ns). This resolution in the frequency domainexactly corresponds to the resolution achievable by the transmissionperiod of T_(BEOB)=16 ms, i.e. 1/T_(BEOB). The number of samples dependson the total number of time slots in the transponder. From this in turnthe scanning frequency can be determined which corresponds to at leasttwice the frequency of the highest sample in the relevant spectrum(taking into consideration the basic delay T₀). With eight time windows(N_(ZF)) with five time slots (N_(ZS)) each and a basic delay T⁰=1000ns, there is a scanning frequency of at least 2*(5*8*25 ns+1000 ns)*2.5Hz/ns=10 kHz. In principle, the entire information contained in theresponse signals is now contained in the spectrum which is divided intofrequency windows and frequency slots completely analog to the timedomain. Decoding is carried out by testing which frequency slot in thecorresponding frequency window is occupied by a response signal.

In order to make things more clear, for the following statements it isagain changed to a description in the time domain.

If a transponder contains an encoding unit without calibrating unit,considerable problems result during practical application. Inparticular, the field of application of the system is limitedconsiderably. The measurement of the propagations T_(A),T_(B), . . .,T_(E), which is necessary for identification, must be carried out soexactly that a clear allocation to a time slot is possible. This iscertainly possible if the precision of the propagation measurementcorresponds to at least half the length of the identifying time slots,e.g. 12.5 ns.

Propagation of the identifying response signals is changed generally (tothe same extent for all) by:

1. changes in the air gap length (reading distance);

2. changes in the length of the antenna supply line;

3. changes in the propagation of the interrogation station, for exampleby aging or temperature changes of electronic components,

4. temperature-dependent changes in the delay T₀-T_(SIT) (T_(SIT) is themostly variable part of the basic delay T₀ lying outside thetransponder) which is the same for all response signals in thetransponder,

to mention only some important general disturbing influences. Forexample, additional 2 meters of antenna cable and 3 meters air gapalready lead to a propagation increase of 40 ns if it is taken intoconsideration that the signal passes to the transponder and back again.

The propagation of the response signal is additionally changedindividually (to a different extent for each response signal) by:

1. temperature-dependent propagation variations in the individualacoustic channels of the encoding unit,

2. changes of the signal speed in the individual acoustic channels ofthe encoding unit,

to mention some important individual disturbing influences.

A calibrating response signal R_(KAL) is used which measures the generaldisturbing influences on the propagations of the response signals. Forthis, according to the invention, exactly one response signal isgenerated whose position in a time slot of the calibrating time windowis exactly known in advance to the interrogation system. T_(KAL) is thepropagation of the calibrating response signal.

A process for reduction of the general disturbing influences isdescribed in the following, which process can, in accordance with theinvention, be used for the above interrogation system with chirpsignals. For this, in turn, it is changed to a description in thefrequency domain.

As a result of the mentioned disturbing influences, in particular thegeneral ones, the spectrum of the response signal is no longer, asdesired, on the discrete sample grid of the FFT. Starting from a certaindegree, the shifts can even lead to a wrong reading of the code. Asmentioned above, according to the invention a calibrating responsesignal is therefore generated whose transponder-generated delay is knownto the interrogation station and equal for all transponders of oneapplication.

This calibrating time window can preferably be arranged on thetransponder as the first time window KAL (cf. FIGS. 2a, b). In otherembodiments the calibrating time window can also preferably be locatedbetween the identifying time windows or even at the end.

With a suitable weighting of the sample values before transformation(so-called FFT windows), a good decorrelation of the individualidentifying responses in the spectrum can be achieved if there is acertain distance between the calibrating time slot (KAL0, ΔT^(KAL)_(ZS)) and the subsequent identifying time slots. This distance isrealized by the calibrating off-time slot (KALS, ΔT^(KAL) _(ZS)). Theexact position of the maximum of the calibrating response signal can bedetermined in the spectrum. For this, only the closest environment ofthe expected position must be checked in order to exclude possibledisturbing signals such as e.g. triple reflexes. This can be done by theknown matched filter technique or by correlation since the desiredposition of the calibrating response signal is known to theinterrogation means. Consequently, with respect to the theoretical,transponder-independent, expected frequency of the calibrating signal,there is an exact value for the frequency error Δω being independent ofthe sample grid.

By subsequently multiplying (mixing) the sampled response signal in thetime domain, wherein the time signal preferably remains stored in thesignal processor, with the complex function

f(t)=e ^(−jΔωt)  (2)

a spectral shift of the time signal by the frequency error Δω isachieved. The now calibrated, complex time signal now undergoes the sameFFT which was used for determining the frequency error. Now the spectraof all response signals lie almost exactly on the desired samples of theFFT, and a decoding error due to non-determined general propagationshifts can thus be excluded.

The interrogation means can subsequently interrogate the spectralsamples of each identifying time slot or frequency slot for itsamplitude. Since due to the used kind of position encoding one frequencyslot per frequency window must be occupied, the maximum sample within anidentifying frequency window corresponds to the desired position of theidentifying response signal.

To avoid decoding errors if there are weak receiving signals, outsidethe frequency range occupied by the response signal, preferably in therange corresponding to the basic delay T₀, the noise level of the entiresignal received by the interrogation station is then determined. If theratio between the detected maximum in the identifying frequency slotsand the noise level is too low, the encoded information is rejected.

The spectrum for evaluating the response signal calibrated in accordancewith the invention also corrects in particular the temperature-dependentshift of the basic delay T₀. The individual part of the propagationshift of the response signals, however, is not corrected. This can beexplained on the basis of the situation exemplarily shown in FIGS. 2a,b. In this embodiment, the calibrating time window (KAL) having thelength ΔT^(KAL) _(ZF) lies in front of the identifying time windows. Theshifts of the basic delay T₀can be corrected in this arrangement.However, for the subsequent identifying response signals (A,B, . . . )the temperature is still effective. At a temperature change of 100° anda temperature coefficient of about 70 ppm/° C. for lithium niobate thereis a relative propagation delay of 0.7%. This shift starts to having anegative effect on decoding if the shift of all identifying time windows(A,B, . . . ) in the time interval T2 reaches half of the time slotduration ΔT_(ZS), e.g. 25 ns/2=12.5 ns. It is an advantageous feature ofthe position encoding according to the present invention that thisindividual disturbing influence does not start to have a negative effectbefore in the preferred embodiment the number of all identifying timeslots is larger than e.g. about 70. In the preferred embodimentcomprising five time slots each, for a temperature range of 100° C. amaximum of about 70/5=14 identifying time windows (A,B, . . . ) or about70×4/5=56 identifying time slots can be used without further correctingthe calibration.

For transponders having more identifying time slots or for a highertemperature range, either the length of the time slots can be increased,or a second calibrating time slot can be introduced, or a furtherfrequency correction δΔω can be determined by means of a gradientalgorithm (calibration correction). For this, the further frequencycorrection δΔω is determined from the theoretically linarly increasingshift of the spectral centers of the calibrated and correctly decodedresponse signals (until and including e.g. time slot K) by correlationwith the desired response signal. According to the invention, the valueδΔω corresponds to the frequency error determined from the gradient,said frequency error corresponding to the temperature-dependentindividual propagation delay for the identifying time slot K being thelast one which was evaluated so far.

In the same way as described above, for the next amount of time slots(I,J, . . . ) of the stored response signals the spectrum of the timesignal is again spectrally shifted by the value δΔω by means of saidcomplex multiplication (equation (2)) in the time domain and againFourier transformed. For said next amount of time slots (I,J, . . . )the temperature-dependent shift is also not disturbing. Thecorresponding identifying response signals can now also be decoded.

For further securing the encoding, the temperature-dependent gradient inthe delay of subsequent time slots can be determined in said secondamount of time slots and compared with the frequency correction value Δωof the first amount. In a further embodiment the signal quality can beimproved by filtering subsequent detection processes (a plurality ofchirps each lasting T_(BEOB)=16 ms).

In sum, there are, for example, the following process steps (cf. FIG.3):

a) storing the response signal as a time signal in the interrogationstation;

b) weighting with a suitable window function;

c) determining the spectrum of the weighted time signal (discreteFourier transformation);

d) determining the position of the calibrating sample in the spectrum(determining Δω);

e) generating a calibrated time signal by multiplying the stored timesignal by exp(−jΔωt);

f) weighting with a suitable window function;

g) calculating the discrete Fourier transformation of the calibratedtime signal;

h) determining the locally maximum sample within each frequency windowwith a maximum of k frequency slots;

i) determining the noise level;

j) interrupting the further evaluation if the signal-to-noise ratio liesbelow a predetermined threshold;

k) storing code part 1;

l) estimating the gradient of the temperature-dependent shift within theselected code part 1;

m) determining the frequency shift of the last (i.e. k-th)

frequency window of code part 1, i.e. the value δΔω;

n) correcting the calibrated time signals by multiplication withexp(−jΔωt);

o) weighting with a suitable window function;

p) determining the discrete Fourier transformation of the correctedcalibrated time signal;

q) search for the locally maximum sample in each frequency window to theremaining (higher) code parts (up to a maximum of 2 k window slots);

r) again checking the S/N ratio;

s) storing the second code part;

t) analog processing of the entire code.

In the following a process for determining the information contained inthe measuring response signals is described, which process can,according to the invention, be used for an interrogation system withchirp signals and transponders which comprise at least one identifyingunit.

According to the invention, in said transponder at least two acousticchannels each having one time window (MESA, MESB) having the lengthT_(MESA), T_(MESB) are used for measuring purposes, in particular formeasuring the temperature of the transponder. (In FIG. 2a T₃ is thepropagation of the measuring response signals). It is advantageous inthis connection if these two channels overlap so that the measuringresponse signals (R_(MESA), R_(MESB)) have a long, common signal line.For example, the process described in the patents of X-Cyte, i.e. theprocess of measuring the phase of the two measuring signals, can be usedfor measuring the temperature. The temperature change can be determinedon the basis of the change in the difference of the two phases. Inprinciple it is known and described in the patents of X-Cyte how thisphase measurement is carried out.

In a preferred embodiment the measuring time windows (MESA, MESB)ΔT^(MES) _(ZF) contain only one time slot and one off-time slot and arearranged after the identifying time slots. The length of thecorresponding time slots (MESA0, . . . MESB0, . . . ) ΔT^(MES) _(ZF) isselected such that there is no angle ambiguity (more than 360°) and asufficiently good sensitivity is achieved. In a preferred embodiment thelength of the measuring time slots (MESA0, MESB0) and the length of thecorresponding off-time slot (MESAS, MESBS) ΔT^(MESS) _(ZS) is e.g. 25ns. With lithium niobate there is thus, for example for a temperaturechange of 100°, a relative phase change of about 306°/100° C. or about3°/° C.

With the above arrangement only the relative temperature can bemeasured. The absolute temperature can be calculated since with thecombination of encoding unit and measuring unit according to theinvention it is possible to store calibrating information belonging tothe corresponding transponder code in the interrogation station.

In a preferred embodiment, the encoding unit 11 and the calibrating unit12.1 are realized as a SAW element (cf. FIG. 4). In the present examplethe encoding unit 11 comprises five encoding channels 11.1 to 11.5 whichgenerate one identifying response signal component each. The encodingchannels are physically realized as five acoustic channels. In any oneof the acoustic channels one transducer electrode 16 and one reflector17 are realized. The reflectors are arranged such that the fivementioned encoding channels 11.1, . . . 11.5 have a code-specificlength.

The delay line 14 being also implemented on a SAW element is locatedbetween the antenna 10 and the encoding unit 11. It makes sure that theresponse signal has the mentioned basic delay T₀. According to theinvention, the basic delay T₀ is not generated in the individualencoding channels 11.1 to 11.5 but in a separate channel being connectedupstream or downstream. If the basic delay is carried out in eachindividual encoding channel (i.e. together with the encoding), a muchlarger substrate surface is required than in the embodiment according tothe present invention. For this it is not urgently necessary that theentire delay T₀ (initial dealy) is realized in the separated signal line14.

The interrogation signal received by the antenna 10 is coupled into theacoustic channel 14.1 of a SAW element via a first transducer electrode16; there it is delayed by a time t_(delay)/2 (<=T₀/2), coupled out viaa second transducer electrode 16 and supplied to the encoding unit 11via an electrical line. In the latter the interrogation signal isreflected and delayed in accordance with the corresponding length of theencoding channels and delayed for a second time in the signal line 14 byt_(delay)/2. The entire signal delay by the signal line 12 is thus 2t_(delay)/2=t_(delay). In order to generate e.g. t_(delay)=1 μs, on a128-LiNbO₃ substrate a length of L=1990 μm is needed, i.e. about 2 mm.Preferably, t_(delay) is about as long as or slightly longer (e.g. 10%)than T₂.

FIG. 4 shows an optional extension of the SAW unit 11, i.e. furthercalibrating units 12 and measuring units 13 having acoustic channels areprovided in addition to the encoding unit 11. These further units are,for example, directly connected with the antenna 10, however, they canalso be led via the delay line 14. In an acoustic channel which isrealized in accordance with the encoding channels and is the same forall transponders of at least one application, the calibrating unitgenerates said calibrating response signal (R_(KAL)). By placing thereflector at different positions in the calibrating channel, agroup-specific encoding can be generated with otherwise the sameencoding by using the position of the calibrating response signal as agroup code. The measuring unit comprises, for example, two channels13.1, 13.2 for generating measuring response signal components (whichcan, in particular, be used for temperature measurement).

An individual propagation-increasing signal line can be introduced intounits 12 and 13. That means that, if it is desired, a transponder canalso have a plurality of common delay lines having a different length.

FIG. 5 shows a preferred embodiment of the invention, wherein in anarrangement corresponding to FIG. 2 an encoding unit and a calibratingunit are realized on only one SAW chip 18. In addition to the fiveencoding channels 11.1 to 11.5 mentioned in this example, there areadditionally provided a channel 14.1 as a propagation-increasing signalline and a calibrating channel. In this embodiment said delay line willpreferably be somewhat smaller than T₀/2 so that the triple reflex fromthis channel 14.1 lies before the calibrating response signal fromchannel 12.1.

In this embodiment the transducer 21.1 emits in both directions, i.e. onthe one hand to an 180° reflector 19 and, on the other hand, to atransducer 21.2. Five transducers 22.1, . . . , 22.5 emitting in onedirection are connected in parallel with the transducer 22.2. They emitsurface waves in the direction of the code-specifically positionedreflectors 23.1, . . . , 23.5.

The advantage of the geometrical arrangement according to the preferredembodiment lies in that the electrical connecting lines (busbars)between the transducer electrodes 21.2, 22.1 to 22.5 lying in one lineat the edge of the chip are relatively short and thus resistance lossescan be minimized. The connecting lines shown in FIG. 5 should have asmall resistance in the sense of the following criterion:

2πf*R*C<<1,  (3)

R is the parasitic line resistance, C is the entire static capacity ofthe IDT electrode structure (IDT=interdigital transducer) and f is theoperating frequency (e.g. f=2.45 GHz).

According to the invention, in a further embodiment of the combinationof encoding unit 11, calibrating unit 12 and delay line 14 thedisturbing internal reflections on the transducers 21.2, 22.1 to 22.5are considerably reduced visa-vis the useful signal components (e.g.more than −20 dB) by the measures described below (in particular 90°reflectors and RMSC structures).

FIG. 6a shows a variant of FIG. 5 in which the coupling, which is shownin an embodiment comprising three encoding channels, between theacoustic channels 14.1, 11.1 to 11.3 (FIG. 6a) is not carried out bymeans of transducer electrodes 22.2 and 22.1 to 22.5 (FIG. 5) but bymeans of correspondingly arranged 90° reflectors 25, 26.1 to 26.3. Thechannel 14.1, which is arranged on the top in the embodiment accordingto FIG. 6a, comprises a 90° reflector 25 which reflects the surfacewaves out of the channel in an angle of 90° (in the embodiment accordingto FIG. 6a to the “bottom”). The 90° reflectors 26.1 to 26.3 are alignedsuch with respect to the 90° reflector 25 that part of the surface wavecoupled out in the 90° angle is coupled into the corresponding channel11.1 to 11.3. There it passes to the reflectors 23.1 to 23.3 (180°reflectors), is reflected and passed back into the channel 14.1 via the90° reflectors.

The angle between the inclined finger electrodes of the 90° reflectorsand the crystallographic x-coordinate is close to 45° and depends on theanisotropy of the substrate used. (With respect to the dimension of such90° reflectors—which is known in a different connection—it is referredto the book “Surface-wave devices for signal processing”, D. Morgan,Elsevier, 1985).

It is a matter of fact that at least the 90° reflectors 26.1, 26.2 mustbe semi-transparent.

A variant would result if the encoding channel 11.1 and the calibratingchannel 12.1 (FIG. 6a) were caused to partially overlap by inserting asecond semi-transparent reflector 27 into the encoding channel 11.1. Inthis arrangement the reflector 27 generates a calibrating responsesignal and, at the same time, allows passage of part of the incidentsurface waves.

The embodiment according to FIG. 6a has the following advantages:

1. In case of inclined reflectors, the width of the electrode fingers isenlarged, e.g. with 90° reflectors by the factor 2 larger than thecorresponding width with 180° reflectors (vertical wave incidence). Thisis an important advantage for the 2.45 GHz frequency range.

2. The resistance losses in the electrodes are lower since there is noelectric current flow from one electrode to the other (all currents arelocal).

3. There are almost no disturbing internal reflections; thereby inparticular the disturbing triple reflections (between the reflectors andtransducers in FIG. 5) are omitted.

4. The wave propagation transversely with respect to the x-coordinate ofthe crystal introduces additional signal delays which are preferred inthe invention. The chip surface can be optimized or minimizedaccordingly.

In sum, what makes this embodiment stand out is thus a greatdimensioning flexibility, a manufacturability which is nowadayssimplified (higher yield) and considerably reduced internal disturbingreflexes.

FIG. 6b shows a further possibility of coupling the propagation-delayedchannel 14.1 with encoding or calibrating channels, for example threeencoding channels 11.1 to 11.3. Coupling is realized by a RMSC structure28 (reversing multistrip coupler) as exemplarily shown in FIG. 6b.

This structure is an electrode structure which is known from otherapplications and which is surrounded in the present embodiment by aring-shaped (or C-shaped) electrode 29 (ring ground electrode) in orderto be able to avoid technically involved and thus expensive electrodecrossings. Further explanations of the known RMSC structure aresuperfluous (cf., e.g., E. Danicki, “A SAW resonator filter explaitingRMSCs”, 1994, Frequency Control Symp. Proc.).

The RMSC structure can operate on the basic frequency and the secondharmonic. At an operation in the basic frequency the width of theelectrodes is λ/6 (e.g. λ/6=0.26 μm at a basis frequency of 2.45 GHz).At an operation in the second harmonic the electrode widthadvantageously doubles (e.g. 0.52 μm).

RMSC structures have numerous advantages for realizing, for example,encoding channel and calibrating channel having a common delay line 14on one SAW chip in accordance with the invention:

1. The known electromechanical coupling of the surface waves leads tothe fact that a disturbing back reflection in the same channel is veryweak. In particular, the disturbing reflections on the transducer 21.2and multiple reflections in the encoding channels 11.1 to 11.5 (FIG. 5)can be reduced to a very large extent.

2. Furthermore, wave transmission and reflection of one channel into theother is almost complete if the number of electrode fingers issufficiently large, i.e. has a dimension of 3*(ΔV/V)⁻¹, wherein ΔV/V isthe known coupling factor for describing the strength of the piezoeffect in surface acoustic waves.

The embodiment according to FIG. 6b thus uses a RMSC structure. Theelectrode 29 surrounds all four channels 14.1, 11.1 to 11.3. The waveincident into the RMSC structure in the uppermost channel 14.1 is, forexample, coupled into the three channels 11.1 to 11.3 and emitted tohave a reversed propagation direction. Correspondingly, from thechannels 11.1 to 11.3 the code-specific reflections are returned intothe channel 14.1 and emitted there in the opposite direction. Accordingto a particularly preferred embodiment, the electrode fingers aredeformed to have a V-shape between the encoding channels 11.1 to 11.3.It is thus avoided that the surface waves are emitted between theencoding channels 11.1 to 11.3. Instead of the V-shaped deformationsalso other deformations can certainly be realized (for preventingundesired SAW radiation).

It would also be possible to replace the reflectors 23.1 to 23.3 withoutput transducers both in FIG. 5 and FIGS. 6a and 6 b so that in theindividual encoding channels 11.1 to 11.3 incident interrogation signalsare not reflected and passed to the antenna 10 via the transducer 21.1,24, or 30 but that the response signals generated in the encodingchannels 11.1 to 11.3 are passed directly to the antenna 10 via saidoutput transducers.

128-LiNbO₃ is preferably used as the substrate for the SAW element. Whatmakes this material stand out is the strong piezoelectric coupling whichis important for preventing the losses of relatively wide-bandedtransducers. Furthermore, this material has a high surface wave velocityand a low level of parasitic bulk waves. However, other substrates canalso be used. For the embodiment comprising the 90° reflector, forexample, YZ-LiNbO₃ is advantageous due to the low diffraction losses.

The transducers in all embodiments can be realized as so-called splitelectrode transducers operating on the third harmonic. In this case, thewidth of the electrode for 128-LiNbO₃ is about 0.6 μm (like thedistances between the electrode fingers). For so-called self matchedtransducers (i.e. transducers for which the imaginary part of admittanceis close to zero) a structure with about 25 split electrodes (50electrode fingers) is necessary. The real part of admittance is close to50 Ohms if the aperture is close to 100 λ (λ=wavelength of the surfacewave, e.g. 1.6 μm at 2.45 GHz). The duration of the pulse response ofsuch a transducer is about 15 ns.

The reflectors can operate either at the basic frequency or at the thirdharmonic. In the first case about 40 to 50 electrode fingers arenecessary for a strong total reflection. This means a time delay withinthe reflectors of about 16 ns. Since the production technologicaldemands on electrodes having a width of 0.4 μm are high, often splitelectrodes are use which operate on the third harmonic and which have awidth of about 0.6 μm for LiNbO₃ at 2.45 GHz. The number of splitelectrodes lies typically in the range of 15 to 25.

Reflector structures which, according to the invention, are suitable forthe use in transponders (cf. FIGS. 6d and 3) are periodical(period=pitch) and consist of electrode fingers 31, gaps 32, andelectrical connections between the fingers and operate on the secondharmonic. The width w of the fingers is preferably about half theperiod, w=2*λ/4, wherein λ is the wavelength at the operating frequencyν, wherein ν=2.45 GHz.

In a preferred embodiment the second harmonic reflectors are realized ona 128°-LiNbO₃ substrate material with a metal layer thickness of about800 Å.

FIG. 6d describes a preferred embodiment of a second harmonic reflectorhaving six electrode fingers and five gaps 32. The number of electrodefingers is variable and can be at least 1 and at most about 50. Thewidth of the electrode fingers, e.g. 31.1, is preferably the same as thewidth of the gaps 32. The width w of the electrode fingers on128°-LiNbO₃ is about w=2*λ/4=0.78 μm, wherein λ is the wavelength of theSAW signal at an operating frequency of 2.45 GHz. Via a connection atthe upper end 33.1 and lower end 33.2 of the finger, all electrodefingers are electrically connected with each other. The width of theconnection is preferably about 10 μm; however, a smaller or larger widthcan also be selected. The preferred width of the reflector or the lengthof the electrode finger is between about 150 to 200 μm; however, othervalues are also possible.

The reflectivity R (in dB) of said reflector at a center frequency canbe calculated by R (in dB)=20*log(tanhyp(N*r)), wherein n is the numberof electrode fingers and r is the reflection coefficient of theindividual finger. As compared to third harmonic reflectors on a128°-LiNbO₃ substrate material whose metal layer has a thickness of 800Å and N=12 fingers, with second harmonic reflectors a reflectivity canbe achieved which is 6 dB higher.

FIG. 6e describes an embodiment according to the invention of a secondharmonic reflector with a different kind of electrode finger connection.Two fingers each, e.g. 31.1 and 31.2 are connected to a couple 34. Thewidth of the gaps 32 and that of the electrode fingers is almost thesame. The width of the connection (35.1 and 35.2) is not critical, butabout 10 μm is preferred; however, a larger or smaller width can also beselected. Further kinds of electrode finger connections are possible.

FIG. 6f shows a preferred arrangement of three offset reflectors 39.1,39.2, 39.3.

An essential feature of the offset reflectors is that they can bedivided into two approximately equal halves 36.1 and 36.2, 37.1 and27.2, 38.1 and 38.2 as is shown in FIG. 6f, wherein one half 36.2, 37.2,38.2 is slightly displaced with respect to the corresponding other half36.1, 37.1, 38.1.

The amount of displacement depends on the value of a fraction of thewavelength λ. When selecting the amount of displacement it must be takeninto account that if this amount is selected unfavorably thereflectivity of the offset reflector can strongly decrease for theusable signals although both halves have a high reflectivity when beingconsidered alone. The amount of displacement as well as its directioncan be different for different offset reflectors in the same layout. Theright combination of these two influencing values is a particularlyimportant feature.

An incident SAW signal 40 is divided by the offset reflector 39.1 intotwo signal parts 41.1 and 41.2. Part of the incident signal 40 isreflected by each half and forms a first useful signal, a further partis transmitted and reflected on the subsequent reflector 39.2. Part ofthe signal reflected on the reflector 39.2 is again reflected on thereflector 39.1 and leads to a disturbing multiple reflection; the otherpart is transmitted and forms a second useful signal.

In the present arrangement the two transmitted signal parts 41.1 and41.2 have to pass distances having different lengths until they reachthe reflector 39.2. In accordance with the invention, this distancedifference is selected such that with the simple round trip reflection42 (transmission by 39.1 reflection on 39.2 reflection on 39.1.reflection on 39.2 transmission by 39.1) a pitch difference of about λ/2is generated between the passing-back partial signals 43.1 and 43.2. Ifthese two partial signals meet the transducer 44, the pitch differencecorresponds to a 180° phase shift of the two simultaneously arriving SAWpartial signals. This causes a strong attenuation of the simple roundtrip disturbing signal emitted via the transponder antenna 10.

In the present arrangement an effective attenuation for the disturbingmultiple reflections 45 and 46 can additionally be achieved. 45 is thuscaused by the reflection on 39.1, transmission through 44, reflection on39.2, and 46 is caused by the reflection on 39.1, reflection on 44,reflection on 39.1. The condition for an effective attenuation is thatthe partial signals generated by the offset reflectors, e.g. 47.1 and47.2 of the multiple reflection 45, have a pitch difference of aboutλ/2. The same applies to the multiple reflection 46. The electricalexcitation at the transducer 44 is considerably attenuated by the twosimultaneously arriving partial signals of the multiple reflections 45and 46, which partial signals, however, are 180° phase-shifted inaccordance with the pitch difference of λ/2.

As mentioned above, FIG. 6f describes a preferred arrangement of threeoffset reflectors, wherein the offset reflectors 39.1, 39.2, 39.3 areeach generated by second harmonic reflectors. The three reflectors areeach divided into two halves, for example reflector 39.1 is divided into36.1 and 36.2. The lower half 36.2 of reflector 39.1 is displaced by theamount Δ1, the lower half 37.2 of reflector 39.2 by the amount Δ2, andthe lower half 38.2 of reflector 39.3 by the amount Δ3. Thedisplacements Δ1, Δ2, and Δ3 can take on different amount anddisplacement directions.

The halves 36.2 and 37.2 are preferably not displaced in the samedirection but in opposite directions. The amount of displacement is alsonot the same for parts 36.2 and 37.2.

According to the invention there are the following conditions for thedisplacements Δ1, Δ2, and Δ3, which conditions can optionally be mettogether or alone:

|4*Δ2−2*Δ1|=λ/2  (A)

|4*Δ1=λ/2  (B)

 |2*Δ1−2^(*Δ3|=λ/)2  (C)

In accordance with the present invention, displacements in the directionof the non-reflected incident SAW signal 40 are counted positively anddisplacements in the opposite direction negatively. The conditionsmentioned in equations (A-C) correspond to the ideal case for an exactdistance difference of λ/2. However, even if there is an error of up toabout 10% from λ/2, the disturbing influences are still attenuatedadvantageously.

Further conditions can also be formulated which, however, are mostly notvery important for the use in transponders.

A preferably advantageous solution of equations (A-C) is achieved if thefollowing displacements are selected:

Δ1=λ/8

Δ2=−λ/16

Δ3=−λ/8

In the above embodiment displacement is thus performed such that part36.2 is displaced by λ/8 (Δ1=λ/8) in the direction of the incident SAWsignal 40 and part 37.2 is displaced by λ/16 (Δ2=−λ/16) in the oppositedirection. For the simple round trip reflection there is, according toequation (A), a pitch difference between the two partial signals of λ/2.

By selecting Δ1=λ/8 the multiple reflection 46 is additionallyattenuated to a large extent and by selecting Δ3=−λ/8 also the multiplereflection 45.

As a consequence of the arrangement according to the invention anadditional attenuation of about 10 dB can be achieved for the disturbingreflections.

There are a number of known possibilities for realizing the transducer44. At an operating frequency of 2.45 GHz a preferred variant is the useof an interdigital transducer with split electrode fingers whichoperates on the third harmonic. For a 128°-LiNbO₃ substrate materialthere is an advantageous embodiment with about 25 split electrodefingers and a finger length of about 150 μm to 200 μm. Under thiscircumstances the transducer impedance is about 50 Ohms and anadditional adaptation is not required.

The different embodiments can be combined with each other or can bemodified. The different ways of coupling the acoustic channels can inprinciple be combined. Acoustic channels can be connected in paralleland in series. The arrangement of the different channel is preferablyselected such that there are minimum losses, minimum space requirementsand minimum internal disturbing reflexes.

What is claimed is:
 1. A surface acoustic wave (SAW) element having apiezoelectric substrate for launching a surface acoustic wave into anacoustic channel, said piezoelectric surface comprising at least oneinterdigital transducer (IDT) arranged in a first direction on thesurface of the substrate and a reflector arranged on the same surface inan interior of the acoustic channel and oriented such that the reflectorreflects the surface acoustic wave in a second direction opposite to thefirst direction, and wherein the reflector is divided laterally withrespect to the first and second directions into at least two areas whicheach open up a subchannel, and at least one of said two areas is offsetwith respect to the direction of the acoustic channel.
 2. A SAW elementaccording to claim 1, wherein the offset is about λ/8, and wherein λ isa wavelength generated by the interdigital transducer (IDT) at a centeroperating frequency of the SAW element.
 3. A SAW element according toclaim 1, wherein the first and second areas have equal apertures.
 4. ASAW element according to claim 1, wherein a double reflected echo signalis compensated due to an additional acoustic path |λ/8*4|=λ/2 and acorresponding phase shift of 180° between waves in the subchannel.
 5. ASAW element according to claim 1, wherein the reflector is a calibrationreflector.
 6. A SAW element according to claim 1, wherein the transduceris bidirectional and is able to emit surface acoustic waves with sameamplitudes to opposite directions.
 7. A SAW element according to claim6, wherein at least one further reflector is arranged in the acousticchannel at a side opposite to the transducer, wherein the furtherreflector is divided laterally with respect to the first and seconddirections into at least two areas which each open up a subchannel, andat least one of said two areas is offset with respect to the directionof the acoustic channel.
 8. A SAW element according to claim 7, whereinthe reflectors arranged on opposing surfaces of the interdigitaltransducer are divided in a lateral direction into two areas havingequal apertures and one of said areas has an offset in the direction ofthe IDT or away from the IDT which offset is λ/8, wherein λ is awavelength generated by the IDT at the center operating frequency of theSAW element.
 9. A SAW element according to claim 8, wherein a round-tripecho signal generated by one single reflection on two reflectorsarranged on both sides of the transducer is attenuated due to a 180°phase shift between the surface acoustic waves propagating in the twosubchannels.
 10. A SAW element according to claim 7, wherein a secondreflector is arranged in the acoustic channel in a greater distance fromthe IDT than the first reflector, wherein both reflectors are dividedinto two areas having an equal aperture and upper and the lower areas ofthe first reflector are offset by a distance |Δ41| along the acousticchannel away from the transducer (+x direction: offset Δ1=+|Δ1| ortowards the transducer (−x direction: offset Δ1=−|Δ1|), whereas thecorresponding area of the second reflector is offset by the distance|Δ2|.
 11. A SAW element according to claim 10, wherein the offsetΔ1=|Δ1|=λ/8 and Δ2=−|Δ2|=−λ/16.
 12. A SAW element according to claim 10,wherein at least one reflector does not comprise an offset between theareas.
 13. A SAW element according to claim 10, wherein the reflectorsare short-circuited, or open electrodes or electrodes connected ingroups and made of a metal and arranged periodically with periodicdistances of λ/2, λ or multiples of λ/2, and wherein the short-circuitedelectrodes may comprise additional connections.
 14. A SAW elementaccording to claim 13, comprising a plurality of transducers arranged indifferent acoustic channels and electrically connected with each other.15. A SAW element according claim 14, wherein the transducers, thereflectors or both are weighted, by the number of electrodes or theiraperture.
 16. A SAW element according to claim 15, wherein thereflectors are suited for encoding tag data due to their position intime or a time delay value of a reflected response.
 17. A SAW elementaccording to claim 16, wherein all reflectors have an initial delayT_(init) and all reflectors are arranged in a distance from thecorresponding IDT which is greater than a predetermined minimum valueL_(init)=2×T_(init)/v_(SAW), wherein v_(SAW) is the velocity of thesurface wave.
 18. A SAW element according to claim 17, wherein in aplurality of the acoustic channels formed by the transducers one or morereflectors having a transmittance of more than 10% are placed and thereflectors generate response signals having exactly the same delay. 19.A SAW element according to claim 18, wherein at an end of an acousticchannel and close to an edge of the piezoelectric substrate there ismeans for preventing parasitic reverse reflections of the SAW elementthrough the edge of the substrate, by absorption, scattering or inclinedreflection of the wave.
 20. A SAW element according to claim 13 whereinthe electrodes are made of a metal selected from the group consisting ofAl and Au and the reflectors are divided laterally perpendicular to theacoustic channel.